Switched resonant ultrasonic power amplifier system

ABSTRACT

A switched resonant power amplifier system for ultrasonic transducers is disclosed. The system includes an amplifier that receives and processes a driver output signal for generating a drive signal that is provided to an ultrasonic device for controlling output of the ultrasonic device. An output control circuit receives and processes a signal related to a feedback signal generated by the ultrasonic device and a divider reference signal, and generates a compensated clock signal that is adjusted for at least one of phase and frequency differences between the received feedback signal and the divider reference signal. A compensated drive circuit receives and processes the compensated clock signal for generating the divider reference signal, and for generating the driver output signal.

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation application of U.S. patentapplication Ser. No. 13/350,877 filed on Jan. 16, 2012, now U.S. Pat.No. 8,485,993, which is a continuation application of U.S. patentapplication Ser. No. 12/163,408 filed on Jun. 27, 2008, now U.S. Pat.No. 8,096,961, which is a continuation application of U.S. patentapplication Ser. No. 10/974,332 filed on Oct. 27, 2004, now U.S. Pat.No. 7,396,336 which claims the benefit of priority to U.S. ProvisionalPatent Application Ser. No. 60/538,202 filed on Jan. 22, 2004, U.S.Provisional Patent Application Ser. No. 60/527,812 filed on Dec. 8,2003, and U.S. Provisional Patent Application Ser. No. 60/515,826 filedon Oct. 30, 2003, the entire contents of which is incorporated byreference herein.

BACKGROUND OF THE INVENTION

1. Technical Field

The present disclosure relates to devices for amplifying an input signaland providing an output signal to a surgical instrument. Moreparticularly, the present disclosure relates to a switched resonantultrasonic power amplifier system for surgical instruments.

2. Background of Related Art

Conventional power amplifier circuits for supplying drive signals toultrasonic transducers are susceptible to drift and droop in powerdelivery and variations in frequency when the ultrasonic transducer isexposed to changing loading conditions. Additionally, conventional poweramplifier circuits require a relatively large footprint, are notlightweight, have efficiency problems, are generally complex circuits,and require heat sinking to dissipate heat generated during operation.Due to their relatively large size and radiated heat, placement ofconventional power amplifier circuits may be problematic in a medicaltreatment facility. Therefore, a need exists for a power amplifiercircuit to supply a drive signal to an ultrasonic transducer and whichovercomes the problems of conventional power amplifiers.

SUMMARY

A switched resonant ultrasonic power amplifier system that has improvedoperating efficiency is provided. The switched resonant ultrasonic poweramplifier system of the present disclosure has reduced heat generatingcharacteristics and a smaller footprint than conventional poweramplifiers. Furthermore, the switched resonant ultrasonic poweramplifier system includes compensation circuitry for changing tissueloads during system operation, structure for frequency, phase, and gainstabilization and structure for ultrasonic power loss compensation.

The present disclosure relates to a switched resonant ultrasonic poweramplifier system including a switched resonant power amplifier. Thepower amplifier system further includes a wave shaping circuit, afrequency generating and compensating circuit, and a compensated drivecircuit. The switched resonant power amplifier generates a transducerdriver signal for driving an ultrasonic transducer. The wave shapingcircuit includes a zero crossing detector and a comparator. A feedbacksignal from the ultrasonic transducer is generally sinusoidal and isapplied to an input of the zero crossing detector where it istransformed into a square wave. The square wave output of the zerocrossing detector is capacitively coupled to the input of the comparatorto form a reset signal.

The frequency generating and compensating circuit includes a referencetimer and a phase-locked loop. The reset signal is applied to an inputof the reference timer to generate a compensated reference signal havinga substantially identical frequency that is further applied to an inputof the phase-locked loop. The phase-locked loop outputs a compensatedclock signal at a particular frequency that is controllable by thecompensated reference signal applied to the input of the phase-lockedloop. The compensated clock signal is generally at a different frequencythan the desired output signal to be applied to the ultrasonictransducer.

The phase locked loop compares the compensated reference signal to adivider reference signal for generating a frequency error signal and/ora phase error signal. The phase locked loop provides frequencycompensation by adjusting the compensated clock signal according to avalue of the frequency error signal. In addition, it may include a phasedelay circuit for adjusting the phase relationship between thecompensated reference signal and the divider reference signal accordingto a value of the phase error signal. Generally, the phase locked loopreceives digital input signals from the drive circuit and the waveshaping circuit. Alternatively, the phase locked loop may be configuredand adapted for mixed-mode signal processing where the inputs are acombination of analog and digital signals. By advantageously adjustingthe compensated clock signal for frequency and/or phase, the ultrasonicpower amplifier system compensates the gain of the ultrasonic amplifiersystem.

The compensated clock signal is applied to an input of the compensateddrive circuit. The compensated drive circuit includes a divider, aflip-flop, and a driver. A selected step-down ratio is applied to thecompensated clock signal in the divider that results in a counter outputsignal delivered by the divider to the flip-flop, which has a lowerfrequency than the compensated clock signal. The counter output signalhas a frequency that is approximately double the selected operatingfrequency for the ultrasonic transducer. A further reduction infrequency occurs as the counter output signal is applied to theflip-flop. The flip-flop generates two complementary square waves thatare substantially 180° out-of-phase with respect to each other. Each ofthe square waves has a frequency that is at the selected operatingfrequency for the power amplifier and approximately one-half of thefrequency of the counter output signal. These complementary square wavesare applied to inputs of the driver for amplification and transmissionto the inputs of the switched resonant power amplifier as driver outputsignals.

In another preferred embodiment, the driver includes a phase delaycircuit that cooperates with the driver and provides phase compensationfor the switched resonant power amplifier input signals. By controllingthe phase relationship between the input signals, the driver is nowphase correlated and random phase relationships are significantlyminimized.

The switched resonant power amplifier includes a pair of insulated gatebi-polar transistors that receive the driver output signals. Theinsulated gate bi-polar transistors are biased such that when one isconducting the other one is not conducting, since one driver outputsignal has a value that corresponds to a “high” value, while thecomplementary driver output signal has a value that corresponds to a“low” value. When the driver signals change states (e.g., high to lowand low to high), the respective insulated gate bi-polar transistorschange from a conducting state to a non-conducting state, therebyproviding an output to a primary side of an output transformer. On asecondary side of the output transformer is a pair of DC blocking outputcapacitors further coupled to an input of an ultrasonic device. Thewaveforms on the primary side of the output transformer are coupledacross to a secondary side of the output transformer, where thewaveforms combine to form the transducer driver signal. The ultrasonicdevice includes an ultrasonic transducer and a feedback transducer thatare operatively coupled to the secondary side of the output transformer.The ultrasonic transducer receives the transducer drive signal from theoutput transformer and drives the transducer element to deliver theultrasonic energy. The feedback transducer generates the feedback signalthat is coupled to the wave shaping circuit.

In addition, the ultrasonic power amplifier system includes an outputcontrol circuit. The output control circuit includes the frequencygenerating and compensating circuit and the drive circuit. It cooperateswith the wave shaping circuit for real time monitoring and control. Thereset signal, that is representative of the feedback signal, is receivedby the frequency generating and compensating circuit for generating acompensated clock circuit. The divider reference signal is compared tothe compensated reference signal in real time to control the compensatedclock signal for frequency, phase, and/or gain. Additionally, the drivecircuit includes a phase delay drive disposed in the driver foradditional phase compensation between switched resonant power amplifierinput signals. By providing real time monitoring and control of thedrive signal to the ultrasonic device, the ultrasonic power amplifiersystem is capable of automatically monitoring and controlling the outputof the ultrasonic device.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the presently disclosed switched resonant ultrasonicpower amplifier system are described herein with reference to thedrawings, wherein:

FIG. 1 is block diagram of a switched resonant ultrasonic poweramplifier system in accordance with an embodiment of the presentdisclosure; and

FIG. 2 is a schematic diagram of an embodiment of a switched resonantpower amplifier of FIG. 1 in accordance with the present disclosure.

DETAILED DESCRIPTION

Embodiments of the presently disclosed switched resonant ultrasonicpower amplifier system will now be described in detail with reference tothe drawings, in which like reference numerals and characters designateidentical or corresponding elements in each of the drawings.

As mentioned above, conventional power amplifier circuits, which supplydrive signals to ultrasonic transducers, are typically susceptible toso-called “drift” and “droop” in power delivery and variations infrequency when the ultrasonic transducer is exposed to changing loadingconditions. Moreover, conventional power amplifier circuits aretypically very complex (e.g., complex circuitry), require a relativelylarge footprint and are quite burdensome, suffer from efficiencyproblems, and require a heat sink (or other cooling means) to dissipateheat generated during operation. As a result, placement of conventionalpower amplifier circuits may be problematic in a medical treatmentfacility.

Referring to FIG. 1, an exemplary embodiment of the presently disclosedswitched resonant ultrasonic power amplifier system 10 is illustrated.Switched resonant ultrasonic power amplifier system 10 is enclosed bybox 12 in FIG. 1 and includes a switched resonant power amplifier 100, awave shaping circuit 125 having a zero crossing detector 130 and acomparator 140, and a frequency generating and compensating circuit 157having a reference timer 150 and a phase locked loop (“PLL”) 160. Theswitched resonant ultrasonic power amplifier system 10 further includesa compensated drive circuit 193 having a divider 170, a flip-flop 180,and a driver 190. An ultrasonic device 200 includes an ultrasonictransducer 114 and a feedback transducer 118 (as shown in FIG. 2) forreceiving a transducer driver signal 116 that is an output of theswitched resonant power amplifier 100. Preferably, driver signal 116 isapplied to ultrasonic transducer 114. A feedback signal 120 is generatedby the feedback transducer 118 and is communicated to zero crossingdetector 130. Feedback signal 120 is proportional to driver signal 116with substantially similar phase and frequency values and generallylower voltage values.

As shown in FIG. 2, switched resonant power amplifier 100 includes aplurality of switching elements 102A, 102B; a corresponding number ofresonant tuning components or elements including a tuning capacitor104A, 104B and a tuning inductor 106A, 106B; and an output transformer108. Tuning capacitors 104A, 104B and tuning inductors 106A, 106B formfirst and second tuning circuits 109A, 109B respectively. Outputtransformer 108 is operatively coupled to an input of ultrasonictransducer 114. A variety of devices may be used for switching elements102A, 102B, including relays, metal oxide semiconductor field effecttransistors (“MOSFET”), and insulated gate bipolar transistors (“IGBT”).

In operation, driver 190 provides at least one driver output signal 195that is coupled to the input of at least one switching element 102.Driver output signal 195 includes a corresponding number of inputsignals 195A, 195B to the number of switching elements 102A, 102B ofswitched resonant power amplifier 100. Each switching element 102A, 102Bis capable of producing an amplified output of the respective inputsignals 195A, 195B. A supply voltage V_(DC) is supplied through tuninginductors 106A, 106B to switching elements 102A, 102B where tuninginductors 106A, 106B are connected in a series relationship to a supplylead of each switching element 102A, 102B. Tuning capacitors 104A, 104Bare connected in a parallel relationship to an output lead of eachswitching element 102A, 102B.

The amplified output of each switching element 102A, 102B is coupled tothe corresponding tuning circuit 109A, 109B. Tuning capacitors 104A,104B and tuning inductors 106A, 106B are selected to correspond to aparticular resonant frequency of input signals 195A, 195B. For example,if the selected transducer driver signal 116 has a frequency of 23 KHz,i.e., a period of 43.5 μs, then the tuned period for each switchingelement 102A, 102B is 21.75 μs. The tuned period for tuning circuits109A, 109B is defined by the formula T=π(LC)^(1/2), where L is the valueof tuning inductors 106A, 106B, C is the value of tuning capacitors104A, 104B, and T is the tuned period.

Output transformer 108, in cooperation with output capacitors 110couples the amplified output of switching elements 102A, 102B, or driversignal 116, to ultrasonic transducer 114. Output capacitors 110 areconnected in a series arrangement with the secondary coil of outputtransformer 108. Using output capacitors 110 in a series arrangementsubstantially blocks any residual direct current (“DC”) and passessubstantially all the alternating current (“AC”) on the secondary sideof output transformer 108. Preferably, output transformer 108 has aratio of approximately 1:1 while output capacitors 110 have a value ofapproximately 10 μf.

In a preferred embodiment, a pair of IGBTs, used as switching elements102A, 102B, is disposed in switched resonant power amplifier 100. Driver190 provides the pair of input signals 195A, 195B that are coupled tothe gates of switching elements 102A, 102B. Input signals 195A, 195B aresquare waves that are approximately 180° out of phase with respect toeach other. Supply voltage V_(DC) is applied to the drains, orcollectors, of switching elements 102A, 102B through series connectedtuning inductors 106A, 106B. Tuning capacitors 104A, 104B areadditionally connected in parallel to the drains, or collectors, therebydefining first and second tuning circuits 109A, 109B. Switching elements102A, 102B further include sources, or emitters, that are connected to achassis common. As each input signal 195A, 195B changes in value, acorresponding inverse change in the output of switching elements 102A,102B occurs.

Each switching element 102A, 102B only conducts when each correspondinginput signal 195A, 195B rises above a threshold value. Using a pair ofswitching elements 102A, 102B permits a first switching element 102A toconduct (e.g., a first input signal 195A is above the threshold value)while a second switching element 102B does not conduct (e.g., a secondinput signal 195B is at or below the threshold value), since thecorresponding first and second input signals 195A, 195B areapproximately 180° out of phase with respect to each other. After aperiod of time, corresponding to the period of first input signal 195A,has elapsed, first input signal 195A is now at or below the thresholdvalue while second input signal 195B is above the threshold value. Atthis point, first switching element 102A stops conducting while secondswitching element 102B begins conducting, thereby providing a switchingcapability of switched resonant power amplifier 100.

Further still, each tuning circuit 109A, 109B is operatively coupled tothe primary side of output transformer 108 and connected in a seriesrelationship to the other tuning circuit 109B, 109A respectively.Selecting the values of L and C, for tuning inductors 106A, 106B andtuning capacitors 104A, 104B, respectively, determines the resonantfrequency of first and second tuning circuits 109A, 109B, respectively.

In an exemplary embodiment, the resonant frequency of each tuningcircuit 109A, 109B is tuned near to the operating frequency of eachinput signal 195A, 195B. When first switching element 102A isconducting, it generates a first output that is operatively coupledthrough first tuning circuit 109A. The output of first switching element102A and its associated first tuning circuit 109A is operatively coupledto the primary side of output transformer 108 and is preferably an AChalf sine wave.

Operation of second switching element 102B and tuning circuit 109B issubstantially similar to the operation of first switching element 102Aand first tuning circuit 109A as described above. Second switchingelement 102B does not conduct when first switching element 102Aconducts, since input signal 195B is approximately 180° out of phasewith respect to input signal 195A. Therefore, the output of switchingelement 102B is essentially an AC half sine waveform that iscomplementary to the output of switching element 102A and provides asubstantially smooth combined sinusoidal output wave at the secondaryside of output transformer 108. The output wave has a frequency that issubstantially equal to the input frequency of input signals 195A, 195B.

Output transformer 108 is preferably configured for a 1:1 primary tosecondary ratio where the output waveform is substantially equivalent inmagnitude to the input waveform. Output capacitors 110 are connected tothe secondary side of output transformer 108 and generally block any DCcomponent of the output waveform that may be present on the secondaryside of output transformer 108. In addition, output capacitors 110conduct substantially the entire AC component of the output waveform,thereby contributing to the smooth sinusoidal AC output waveform. Thedownstream side of output capacitors 110 is connected to the ultrasonictransducer 114, which could be magnetostrictive, piezoelectric, ortransducer structures as is known in the art.

Ultrasonic device 200 includes feedback transducer 118 for providingfeedback signal 120 to wave shaping circuit 125. Output transformer 108is electrically coupled to ultrasonic device 200 such that electricalpower is delivered to ultrasonic transducer 114 as transducer driversignal 116 and converted to ultrasonic power. Furthermore, switchedresonant power amplifier 100 generates transducer driver signal 116 withthe desired signal characteristics (e.g., wave shape, amplitude, and/orfrequency) and communicates it to an input of ultrasonic device 200. Ina preferred embodiment, transducer driver signal 116 is a substantiallysmooth sinusoidal AC waveform with the desired signal characteristicsfor driving ultrasonic transducer 114.

Feedback transducer 118 is also disposed on the secondary side of outputtransformer 108 and generates feedback signal 120 that is electricallycoupled to zero crossing detector 130. In a preferred embodiment,feedback signal 120 is a sample of transducer driver signal 116 having awaveform with substantially the same frequency and wave shape. Sincefeedback signal 120 and transducer driver signal 116 are coupled withinthe ultrasonic device 200, characteristics of feedback signal 120 arerelated to characteristics of transducer driver signal 116 and reflectchanges in the characteristics of the transducer(s) (e.g., ultrasonictransducer 114 and/or feedback transducer 118) of the ultrasonic device200. For example, if the frequency of transducer driver signal 116increases with a corresponding decrease in its period, feedback signal120 has a corresponding increase it its frequency and substantiallymatches the frequency change of transducer driver signal 116. Changes inother characteristics of transducer driver signal 116 result incorresponding changes to the respective characteristics of feedbacksignal 120.

Zero crossing detector 130, in cooperation with associated circuitry,modifies feedback signal 120 and provides an output that issubstantially a square wave 135. In a preferred embodiment, zerocrossing detector 130 includes a comparison circuit, such as an LM393integrated circuit, having biasing circuitry and a diode coupled to theoutput of the comparison circuit. Preferably, feedback signal 120 iscoupled to the input of the comparison circuit for providing a morestable output square wave 135. As a component of wave shaping circuit125, zero crossing detector 130 receives an analog input signal (e.g.,feedback signal 120) and produces a digital output signal (e.g., squarewave 135).

By applying feedback signal 120 to an appropriate input lead of thecomparison circuit, zero crossing detector 130 generates square wave 135having a waveform representative of feedback signal 120. As feedbacksignal 120 transitions above a predetermined (zero) voltage referencepoint, thereby becoming more positive, the comparison circuit conductsand provides a positive portion of square wave 135. The output will beof substantially constant amplitude as long as feedback signal 120 ismore positive than the zero reference point. When feedback signal 120 isat the zero reference point, there is no difference in voltage on theinput leads of the comparison circuit, thereby causing the comparisoncircuit to stop conducting, and provide a zero output. As a result, theoutput of the comparison circuit rapidly changes from a constantpositive value to zero, thereby providing a substantially instantaneoustransition of the output signal.

Once feedback signal 120 transitions below the zero reference point,thereby becoming more negative, the comparison circuit again conductsand provides a negative portion of square wave 135. Zero crossingdetector 130 is biased and configured to provide a rapid change from theconstant positive amplitude to the constant negative amplitude formingthe leading and trailing edges of square wave 135, such that the edgesare substantially vertical. Feedback signal 120 and square wave 135 havesubstantially identical frequencies, even if their respective amplitudesare different.

Square wave 135 is coupled to comparator 140, where square wave 135 ispreferably capacitively coupled to comparator 140. Comparator 140includes a comparison circuit and is preferably coupled to a capacitorcoupling circuit that generally blocks any DC component of square wave135 from being transmitted from zero crossing detector 130 and transmitssubstantially the entire AC component of square wave 135 to comparator140. In a preferred embodiment, comparator 140 includes an ICcomparator, such as an LM393 along with associated biasing and feedbackcircuitry.

As the amplitude of square wave 135 goes positive past the zero voltagereference point, it biases comparator 140 such that the output ofcomparison circuit goes negative, thereby causing the output ofcomparator 140, a reset signal 145, to become more negative. A portionof reset signal 145 is coupled through the feedback circuitry to anotherinput of the comparison circuit, thereby providing feedback to thecomparison circuit to produce a more stable output (e.g., reset signal145). Preferably, reset signal 145 has a substantially identicalfrequency to square wave 135 with a waveform that is substantially 180°out-of-phase with respect to square wave 135.

Reset signal 145 is communicated to an input of reference timer 150 forcontrolling a timing function of reference timer 150. As reset signal145 drops below a predetermined reset threshold value, it causesreference timer 150 to reset. When reference timer 150 resets, itgenerates a compensated reference signal 155 having a substantiallyidentical frequency to reset signal 145, square wave 135, and feedbacksignal 120. Compensated reference signal 155 does not have the samephase characteristics as reset signal 145, but is essentially 180°out-of-phase with respect to reset signal 145 and feedback signal 120.Consequently, compensated reference signal 155 is substantially in phasewith square wave 135.

In an exemplary embodiment, reference timer 150 includes an IC timer,such as a 555 precision timer, having associated biasing and feedbackcircuitry. Reference timer 150 in cooperation with the biasing circuitryis configured for operation as an astable multivibrator that produces asquare wave output. Frequency and amplitude characteristics of thesquare wave are determined by the biasing circuit and the signal appliedto a reset input of reference timer 150. According to an exemplaryembodiment of the present disclosure, reset signal 145 is applied to areset input of reference timer 150 to produce compensated referencesignal 155. Combining the biasing configuration for the reference timer150 in cooperation with reset signal 145 yields compensated referencesignal 155 that has substantially the same frequency as feedback signal120.

In a preferred embodiment, the 555 precision timer and the associatedbiasing circuitry of reference timer 150 are configured to generatecompensated reference signal 155 that has a frequency lower than theselected operating frequency of switched resonant ultrasonic poweramplifier system 10. More specifically, the 555 precision timer and itsassociated biasing circuitry are configured so that when the frequencyof reset signal 145 is below the frequency of compensated referencesignal 155, the biasing circuitry determines (e.g., controls) thefrequency value of compensated reference signal 155 for providingcompensation. In the situation where reset signal 145 has a higherfrequency value than compensated reference signal 155, reset signal 145acts as a trigger for the 555 precision timer causing a correspondingincrease in the frequency of compensated reference signal 155.

An input of PLL 160 is coupled to an output of reference timer 150 forcommunicating compensated reference signal 155. PLL 160 receivescompensated reference signal 155 and compares it to a divider referencesignal 177. When reference signal 155 and divider reference signal 177have substantially identical frequencies, PLL 160 produces a compensatedclock signal 165 having a set frequency that corresponds to thefrequency of the reference signal 155 and divider reference signal 177.In the situation where compensated reference signal 155 has a higherfrequency than divider reference signal 177, PLL 160 lowers thefrequency of compensated clock signal 165 as described below.Conversely, when compensated reference signal 155 has a lower frequencythan divider reference signal 177, PLL 160 raises the frequency ofcompensated clock signal 165 as described below.

Advantageously, PLL 160 includes an IC PLL, such as a 4046 PLL IC chip,and associated biasing circuitry. In a preferred embodiment using PLL160, compensated reference signal 155 is coupled to a signal input ofthe PLL 160 while divider reference signal 177 is applied to a referenceinput of PLL 160. Compensated clock signal 165 is generated by avoltage-controlled oscillator internal to PLL 160 chip and tuned to anoutput frequency. Internally, the frequencies of compensated referencesignal 155 and divider reference signal 177 are compared to produce afrequency error signal at a phase comparator output of PLL 160.

This frequency error signal is applied to the voltage controlledoscillator input for adjusting the output frequency of the voltagecontrolled oscillator. If compensated reference signal 155 has a greaterfrequency than divider reference signal 177, the frequency error signalapplied to the voltage controlled oscillator causes a decrease in theoutput frequency of compensated clock signal 165. In the situation wherecompensated reference signal 155 has a lower frequency than dividerreference signal 177, the frequency error signal applied to the voltagecontrolled oscillator results in an increase of the output frequency ofcompensated clock signal 165.

While the above embodiment provides frequency compensation forcompensated clock signal 165, it may also be desirable to provide phasecompensation for clock signal 165. Frequency generating and compensatingcircuit 157 receives reset signal 145, which is representative of theoutput of ultrasonic device 200. As in the previous embodiment, resetsignal 145 controls the generation of compensated reference signal 155that has substantially the same phase and frequency as feedback signal120. PLL 160 receives compensated reference signal 155 and compares itto divider reference signal 177, which is representative of compensatedclock signal 165, thereby producing a phase error signal. When the phasedifference between compensated reference signal 155 and dividerreference signal 177 is at a minimum value (e.g., substantiallyin-phase), the phase error signal will have a low or first value. Insituations where the phase difference between the signals is at amaximum value (e.g., substantially out-of-phase), the phase error signalwill have a high or second value. If the phase difference betweencompensated reference signal 155 and divider reference signal 177 isbetween the maximum and minimum values, the phase error signal will havea value between the first and second values that is representative ofthe phase difference between the signals.

The phase error signal cooperates with associated circuitry in PLL 160to adjust the timing of compensated clock signal 165 and thereby itsphase relationship to compensated reference signal 155. Moreparticularly, a delay circuit 162, such as that discussed in detailbelow, is included in PLL 160 to control the timing of compensated clocksignal 165 for adjusting the phase timing of compensated clock signal165 in accordance with the phase error signal. When the phase errorsignal indicates that compensated reference signal 155 does not have thedesired phase relationship to divider reference signal 177, the delaycircuit 162 of PLL 160 adjusts the phase timing of compensated clocksignal 165 to change the phase relationship between them and preferablysynchronize them. Changes to the timing of compensated clock signal 165are reflected in divider reference signal 177 that is operativelycoupled to PLL 160. In preferred embodiments, compensated referencesignal 155 and compensated clock signal 165 are substantially in-phasewith one another, thereby generating a phase error signal having aminimum value.

The PLL 160 may be configured and adapted to process signals that areanalog, digital or a combination thereof. In this configuration, inputsto PLL 160 may be analog signals, digital signals, or a combination ofanalog and digital signals (e.g., mixed-mode). In the previousembodiment, the inputs were digital signals (e.g., compensated referencesignal 155 and divider reference signal 177) that were processed by PLL160. In the mixed-mode configuration, PLL 160 receives an analog inputsignal (e.g., feedback signal 120 directly from ultrasonic device 200)and compares it to an analog or digital reference signal, such asdivider reference signal 177, as in the previous embodiment, forgenerating the frequency error signal and/or the phase error signal andadjusting the compensated clock signal accordingly.

In exemplary embodiments of the present disclosure, frequency generatingand compensating circuit 157 includes frequency and phase compensationas discussed hereinabove. The frequency and phase compensation may beprovided substantially simultaneously. By advantageously providingfrequency and/or phase compensation, ultrasonic power amplifier system10 provides gain compensation for reset signal 145 since the desiredfrequency and/or phase of compensated clock signal 165 is maintainedduring operation of ultrasonic power amplifier system 10. Furthermore,power compensation is provided, such as when adjustment and compensationof frequency, gain and/or phase (preferably frequency, gain and phase)is optimized. In addition, compensation for changing tissue loads isadvantageously provided, since tissue loading changes the “tune”, i.e.,the natural frequency of the transducer system (e.g., ultrasonictransducer 114 and/or feedback transducer 118), which is being adjustedand compensated for by the switched resonant ultrasonic power amplifiersystem 10.

By way of example only, assume that the desired frequency is 23 KHz andcompensated clock signal 165 has a frequency of 1 MHz that is sampledand output from flip-flop 180 as divider reference signal 177. Whendivider reference signal 177 and compensated reference signal 155 havesubstantially matching frequencies, the frequency error signal isessentially zero. Therefore, the voltage controlled oscillator continuesto generate compensated clock signal 165 at a frequency of 1 MHz. Ifcompensated reference signal 155 has a frequency greater than the 23 KHzof divider reference signal 177, then the frequency error signal causesthe voltage-controlled oscillator to decrease the frequency ofcompensated clock signal 165 below 1 MHz. This decreases the frequencyof divider reference signal 177 to match the frequency of compensatedreference signal 155, thereby returning switched resonant ultrasonicpower amplifier system 10 to a state of equilibrium at the desiredfrequency. By using PLL 160 to correct changes in frequency as in theabove-given example, switched resonant ultrasonic power amplifier system10 automatically adjusts in real time for frequency variations due tochanging load conditions, power supply variations, or other frequencyshifting conditions. In a similar manner, PLL 160 automatically adjustsand compensates for phase differences between compensated clock signal165 and divider reference signal 177.

The output of PLL 160, e.g., compensated clock signal 165, is coupled toan input of compensated drive circuit 193, and preferably, to an inputof divider 170 where the frequency of compensated clock signal 165 isstepped-down by divider 170 to a desired counter output signal 175.Divider 170 is configurable, using a plurality of input to outputratios, to step-down compensated clock signal 165 to one of a multitudeof different output frequencies. Therefore, switched resonant ultrasonicpower amplifier system 10 is adaptable for a number of differentapplications, devices or systems using different desired frequencies.

In an exemplary embodiment, divider 170 is a 4059 programmabledivide-by-n counter chip having associated biasing circuitry. A clockinput receives compensated clock signal 165 for processing by divider170. Biasing circuitry for divider 170 establishes the step-down ratiofor divider 170 and reduces the frequency of compensated clock signal165 to a desired frequency for counter output signal 175.

Advantageously, the associated biasing circuitry is operatively coupledfor programming the step-down ratio where the biasing circuitry iscontrollable by software and/or hardware switches. Hardware switchesallow the operator to manually change the step-down ratio of divider 170and adjust for different frequency outputs of switched resonant poweramplifier system 10. Using software switches to control the biasingcircuitry allows remote operation of the step-down ratio and furtherpermits automatic control of the biasing circuitry by associatedcircuitry coupled to switched resonant power amplifier system 10,thereby improving the flexibility and adaptability of switched resonantpower amplifier system 10.

Coupled to the output of divider 170 is flip-flop 180 for splittingcounter output signal 175 into complementary square waves (e.g., eachsquare wave is substantially 180° out-of-phase with respect to the othersquare wave) where each square wave has a frequency that issubstantially one-half of the frequency of counter output signal 175. Aportion or sample of one of the output square waves is diverted to acomparator input of PLL 160 as divider reference signal 177, which isdiscussed above. Preferably, flip-flop 180 is a quadruple D-typeflip-flop with clear, such as a 74HC175 integrated circuit withassociated biasing circuitry.

Flip-flop 180 is biased such that when counter output signal 175 isapplied to a clock input of flip-flop 180, the flip-flop 180 outputs Qand ̂Q, which are substantially 180° out-of-phase with respect to eachother. Additionally, the output ̂Q is coupled to a data input offlip-flop 180 for biasing flip-flop 180. By using ̂Q as the input to thedata input, the outputs Q and ̂Q are toggled by counter output signal175 such that each of the outputs Q and ̂Q are substantially 180°out-of-phase with respect to each other and substantially one-half ofthe input frequency of counter output signal 175. Preferably, the outputQ is sampled as divider reference signal 177 for supplying a frequencycomparison signal to PLL 160 as discussed above.

A driver input signal 185 is the output of flip-flop 180 and is furthercoupled to an input of driver 190. Driver 190 amplifies driver inputsignal 185 to supply driver output signal 195 to switched resonant poweramplifier 100. Preferably, driver 190 is selected for amplifying driverinput signal 185 to match the desired input characteristics for switchedresonant power amplifier 100.

In a preferred embodiment, driver 190 includes a CMOS MOSFET driver suchas the MIC4424 along with associated biasing circuitry. Driver 190 haselectronic characteristics that are preferred for use with the switchingelements 102A, 102B (e.g., IGBTs) of switched resonant power amplifier100. Driver input signal 185 includes the outputs Q and ̂Q that arecoupled to inputs A and B, respectively, of the driver 190 as shown inFIG. 2. Driver 190, in cooperation with its biasing circuitry, amplifiesthe components (Q and ̂Q) of driver input signal 185 and communicatesthe amplified signals to outputs A and B as driver signals. Theamplified signals substantially maintain their frequency and phasecharacteristics during the amplification process. Outputs A and B arecombined to form driver output signal 195 and are coupled to the inputsof switched resonant power amplifier 100 as input signals 195A, 195B.

Additional frequency stability is provided by combining wave shapingcircuit 125 with frequency generating and compensating circuit 157 toprovide a desired frequency and/or phase compensated input signal todriver 190. By advantageously matching driver 190 to switched resonantpower amplifier 100, proper coupling between driver input signal 185 andswitched resonant power amplifier input signals 195A, 195B is obtainedthereby effecting the desired amplification by switched resonant poweramplifier 100.

In another preferred embodiment, driver 190 includes one or morecomponents and/or circuits to form a phase delay circuit 192 as areknown in the art. One such circuit includes two 555 timers (not shown)connected in series and associated biasing components. Alternatively,the 555 timers may be replaced by a 556 timer, which includes two 555timers. Another example of a delay circuit includes two 74121 integratedcircuits and associated biasing components. Preferably, the biasingcircuitry in phase delay circuit 192 includes components that areadjustable by the system and/or the operator for adjusting the phaserelationship between switched resonant power amplifier input signals195A, 195B and/or the pulse widths of the input signals 195A, 195B.Advantageously, the above-mentioned delay circuits are capable ofproducing an output signal that is time delayed with respect to theinput signal. In addition, each of the above-mention circuits is capableof producing an output signal that has a width that is less than,greater than, or equal to the input signal's width.

Phase delay circuit 192 advantageously cooperates with driver 190 forcontrolling the phase relationship between switched resonant poweramplifier input signals 195A, 195B and for controlling their respectivepulse widths. In the previous embodiment, switched resonant poweramplifier input signals 195A, 195B were substantially 180° out-of-phasewith respect to each other. However, by adding phase delay circuit 192to driver 190, the timing and the pulse widths of each of the switchedresonant power amplifier input signals 195A, 195B is controllable. Inpreferred embodiments, the phase relationship between switched resonantpower amplifier input signals 195A and 195B is variable between about 0°to a value about 360°, while the pulse widths of the input signals 195Aand 195B are substantially equal to one another. By adjusting the phaserelationship and the pulse widths, ultrasonic power amplifier system 10regulates an output from ultrasonic device 200 having the desiredcharacteristics for a particular procedure.

When the phase relationship between switched resonant power amplifierinput signals 195A and 195B is modified, drive signal 116 is pulsed andthe ultrasonic power amplifier system 10, in turn, produces a pulsedoutput from ultrasonic device 200 rather than a substantially continuousoutput, where the time delay between the output pulses is proportionalto the phase relationship. The duration of pulses output by ultrasonicdevice 200 is adjustable by changing the pulse widths of input signals195A, 195B. Numerous advantageous combinations of pulse width and phaserelationship may be used in ultrasonic power amplifier system 10depending on the particular procedure.

Additionally, driver 190 in cooperation with phase delay drive 192provides phase correlation between switched resonant power amplifierinput signals 195A, 195B. Since the desired phase relationship isestablished and maintained between the input signals 195A and 195B byphase delay circuit 192, random or undesirable phase relationshipsbetween the input signals is significantly minimized.

Changes in the loading characteristics of transducer driver signal 116caused by changes in the loading of ultrasonic device 200 are fed backto zero crossing detector 130 as changes in feedback signal 120. By wayof example only, if ultrasonic device 200 is rapidly unloaded, itsoperating frequency rises and is reflected as a frequency rise infeedback signal 120. This increase in the operating frequency ofultrasonic device 200 is communicated to feedback transducer 118 with acorresponding frequency increase in feedback signal 120. As discussed indetail hereinabove, as feedback signal 120 increases in frequency, zerocrossing detector 130 generates square wave 135 having a correspondingincrease in frequency. The increased frequency of square wave 135 iscapacitively coupled to comparator 140 for generating reset signal 145that reflects the frequency increase in feedback signal 120. Incooperation with reference timer 150, the increased frequency of resetsignal 145 raises the frequency of compensated reference signal 155 thatis communicated to PLL 160.

An increased frequency input to PLL 160, as evidenced by the increasedfrequency of compensated reference signal 155, causes PLL 160 to raisecompensated clock signal 165. A higher frequency of compensated clocksignal 165 is transferred to an input of divider 170 thereby causing acorresponding increase in the frequency of counter output signal 175that is communicated to flip-flop 180. Output from flip-flop 180 issupplied as driver input signal 185 and as driver reference signal 177,both signals having increased frequency. The resulting increase in thefrequency of driver input signal 185 is applied to driver 190 and raisesthe frequency of driver output signal 195. By raising the frequency ofdriver output signal 195, switched resonant power amplifier 100 producesa higher frequency transducer driver signal 116 in response. Preferably,the higher frequency of transducer driver signal 116 is substantiallyidentical to the frequency of frequency feedback signal 120, therebyreturning power amplifier 10 to a steady-state equilibrium conditionwhere transducer driver signal 116 and feedback signal 120 are at thesubstantially identical frequency.

By actively monitoring the output of ultrasonic device 200 throughfeedback signal 120 and adjusting driver signal 116 in response thereto,ultrasonic power amplifier system 10 automatically adjusts the output ofultrasonic device 200 in response to changes in operating parameters inreal time. More specifically, ultrasonic power amplifier system 10includes an output control circuit 197 that includes frequencygenerating and compensating circuit 157 and drive circuit 193. Outputcontrol circuit 197 receives reset signal 145 and generates switchedresonant power amplifier input signals 195A, 195B having the desiredfrequency, phase, and/or gain compensation as discussed in detail above.

By advantageously selecting and using solid-state and/or semi-conductorcomponents, switched resonant power amplifier system 10 can be made tohave a smaller footprint, or size, than a conventional power amplifiercircuit for a comparable output. In addition, switched resonant poweramplifier system 10 produces less heat and is more efficient than priorart systems due to the use of solid-state and/or semi-conductorcomponents in the system.

Although the illustrative embodiments of the present disclosure havebeen described herein with reference to the accompanying drawings, it isto be understood that the disclosure is not limited to those preciseembodiments, and that various other changes and modifications may beaffected therein by one skilled in the art without departing from thescope or spirit of the disclosure. All such changes and modificationsare intended to be included within the scope of the disclosure.

What is claimed is:
 1. A system for controlling an output of a surgicaldevice, the system comprising: a driver configured to provide a pair ofdriver output signals approximately 180° out of phase with respect toeach other; an amplifier including a pair of switching elementsconfigured to receive and process the pair of driver output signals andproduce a respective amplified driver output signal that drives thesurgical device and controls the output thereof; and a wave shapingcircuit configured to receive a feedback signal from the surgicalinstrument and provide a square wave output having a frequency that issubstantially identical to a frequency of the feedback signal, the waveshaping circuit configured to provide a reset signal having asubstantial identical frequency and 180° out of phase with respect tothe square wave.
 2. The system of claim 1, wherein the reset signal iscommunicated to a compensating circuit for controlling the generation ofa compensated reference signal.
 3. The system of claim 2, wherein thecompensated reference signal is compared to a divider reference signalfor producing error signal, wherein the divider reference signal isrepresentative of a compensated clock signal.
 4. The system of claim 3,wherein, when a phase difference between the compensated referencesignal and the divider signal is at a minimum value, the phase errorsignal will have a first value.
 5. The system of claim 3, wherein, whena phase difference between the compensated reference signal and thedivider signal is at a maximum value, the phase error signal will have asecond value.
 6. The system of claim 3, further including a delaycircuit that is configured to control the timing of the compensatedclock signal.
 7. The system of claim 3, wherein the compensated clocksignal is coupled to an input of an input of divider where a frequencyof compensated clock signal is stepped-down by the divider to a desiredcounter output signal.
 8. The system of claim 7, wherein the divider iscoupled to a flip-flop for splitting the counter output signal intocomplementary square waves.
 9. The system of claim 8, wherein eachsquare wave is substantially 180° out-of-phase with respect to eachother.
 10. The system of claim 9, wherein each square wave has afrequency that is substantially one-half of the frequency of the counteroutput signal.